Sensor and method of operating the sensor

ABSTRACT

The invention relates to a sensor and a method of operating a sensor with includes a plurality of sensor elements ( 10 ), each of which includes a radiation-sensitive conversion element ( 1 ) which generates an electric signal in dependence on the incident radiation, and also with means ( 21  to  26 ) for amplifying the electric signal in each sensor element ( 10 ) and a read-out switching element ( 30 ) in each sensor element ( 10 ) which is connected to a read-out line ( 8 ) in order to read-out the electric signal. In order to provide a sensor in which a high stability of the transfer function and a favorable signal-to-noise ratio are ensured while maintaining a comparatively simple and economical construction, the means for amplifying include a respective source follower transistor ( 21 ) whose gate is connected to the conversion element ( 1 ), whose source is connected on the one side to an active load ( 23 ) and on the other side to one side of a sampling capacitor ( 26 ), the other side of the sampling capacitor ( 26 ) being connected to the read-out line ( 8 ) via the read-out switching element ( 30 ), a respective reset element ( 27 ) being connected to the conversion element ( 1 ) so as to reset the conversion element ( 1 ) to an initial state.

FIELD OF THE INVENTION

The invention relates to a sensor with a plurality of sensor elements,each of which includes a radiation-sensitive conversion element whichgenerates an electric signal in dependence on the incident radiation,and also with means for amplifying the electric signal in each sensorelement and a read-out switching element in each sensor element which isconnected to a read-out line in order to read out the electric signal.The invention also relates to a method of operating such a sensor aswell as to an X-ray examination apparatus which includes an X-ray sourcefor emitting an X-ray beam for irradiating an object so as to form anX-ray image, as well as a detector for generating an electric imagesignal from said X-ray image.

BACKGROUND OF THE INVENTION

Large-surface X-ray detectors are customarily used for X-ray examinationapplications, notably in the medical field; such detectors consist of aplurality of sensor elements. The sensor elements (pixels) as a rule arearranged in rows and columns in a sensor matrix. Preferably, use is madeof the so-called flat dynamic X-ray detectors (FDXD). Such detectors areseen as universal detector components that can be used in a wide varietyof X-ray apparatus.

In contemporary FDXD embodiments, the individual sensor elements (matrixcells) comprise a radiation-sensitive conversion element, having anintrinsic storage capacity, and a switching element for reading out thesignal present on the conversion element or the storage capacitanceafter the irradiation. The FDXD preferably utilize conversion elementsin the form of photodiodes of amorphous silicon and scintillatorelements connected thereto, or alternatively photoconductors, for thedirect conversion of the X-rays into electric charges. In other types ofsensors for other radiation, of course, other conversion elements canalso be used.

Diode switches or transistors, notably TFTs (thin film transistors) ofamorphous silicon are preferably used as read-out switching elements. Inorder to read out the signal, present as a collected charge on theconversion element or the intrinsic storage capacitance thereof, theread-out switching elements are activated and the collected charge isconducted to the relevant read-out line. From there it flows to acharge-sensitive amplifier (CSA). Subsequently, corresponding electronicinformation is applied to a multiplexer which conducts this informationto a data acquisition unit for display on a display device in the formof a monitor.

When such detectors are used, notably in the medical analysis practice,it is desirable to reduce the radiation dose so as to limit the dosewhereto the patient is exposed; consequently, only a very small amountof radiation is incident on the individual sensor elements. As a result,the electric signal in the individual sensor elements is also verysmall. Therefore, the aim is to realize sensors or X-ray detectorshaving an as high as possible signal-to-noise ratio.

A particularly high signal-to-noise ratio and detection of small doses,of course, is also desirable for other radiation-sensitive sensors. Inorder to improve the signal-to-noise ratio, the signal can in principlebe amplified already in the individual matrix cell of the detector.

U.S. Pat. No. 5,825,033 discloses a semiconductor detector for gammarays in which the charge generated in each pixel in the detectormaterial is stored in an integration capacitor of a capacitive feedbackamplifier. This integration takes place for all pixels simultaneously.In a so-called Correlated Double Sample-and-Hold circuit (CDSH) thenoise induced by the resetting of the integration capacitor iseliminated. Subsequent to the CDSH, the individual pixels are connectedto a respective unity gain buffer which is connected to a read-out linecommon to each column. The read-out lines are then combined byappropriate multiplexers. The sensor in this case consists of a matrixwith 48×48 individual pixels.

For amplifier circuits for enhancing the signal-to-noise ratio, thesignal amplification and the noise are customarily the essentialcharacteristics considered for evaluation. For practical operation thereis a further criterion in the form of the stability of the transferfunction. For example, when the signal amplification or an offset valueof the amplifier fluctuates in time, offset and gain artefacts occur inthe imaging detector system; such artefacts can only be corrected partlyand with great effort only. Such fluctuations may be caused by changesof the temperature or other operating conditions as well as be due toaging, radiation damage and/or trapping effects in semiconductors.

The threshold voltage and the transconductance are liable to changesignificantly in time, notably in the frequently used thin filmtransistors (TFTs) of amorphous silicon, which can also be used notablyfor the manufacture of integrated amplifier circuits in a matrix cell;this may degrade the stability of the transfer function.

Therefore, it is an object of the present invention to provide a sensorand a method of operating the sensor wherein a high stability of thetransfer function and an attractive signal-to-noise ratio are ensured bya comparatively simple and economical construction.

SUMMARY OF THE INVENTION

This object is achieved by means of a sensor which is characterized inthat the means for amplifying include a respective source followertransistor whose gate is connected to the conversion element, whosesource is connected an active load and to one side of a samplingcapacitor, the other side of the sampling capacitor being connected tothe read-out line via the read-out switching element, and that arespective reset element is connected to the conversion element in orderto reset the conversion element to an initial state.

The active load ideally constitutes a current source which impresses aconstant channel current on the source follower transistor. Thethreshold voltage of the source follower transistor is thus stabilized;this threshold voltage is strongly dependent on the channel current,notably in the case of TFTs of amorphous silicon. As a result of thestable threshold voltage, the condition for correct operation of thesource follower transistor with adequate stability of the transferfunction is satisfied. Therefore, the source follower transistor has astable voltage amplification of 1. It is converted into a chargeamplification G_(Q)=C_(S)/C_(P) by the sampling capacitor, wherein C_(P)is the capacitance on the conversion element and C_(S) is thecapacitance of the sampling capacitor. The capacitance on the conversionelement may again be an intrinsic storage capacitance of the conversionelement or an additional capacitance.

Preferably, the active load, the read-out switching element and thereset element are also formed by transistors. All components requiredfor the invention can then be integrated directly in the sensor elementswhile using the thin film technology which is used any way to form thesensor elements; in the context of this technology the transistors canbe made of amorphous silicon or polycrystalline silicon. Because of thestable amplification circuit constructed in conformity with theinvention, the use of the TFT transistors of amorphous silicon that canbe economically manufactured is not a drawback.

A process with vertical integration can now be advantageously used insuch a manner that the surface area of the conversion element, or thestorage capacitance within a sensor element, is not reduced.

In one embodiment a discharge switching element, preferably in the formof a transistor, for example a TFT of amorphous or polycrystallinesilicon, is connected parallel to the sampling capacitor. This dischargeswitching element can be used for the simultaneous, accelerateddischarging of the sampling capacitor during a reset of the conversionelement by means of the reset element, so that the sampling capacitor isalso reset to an initial state.

The reset element and the discharge switching element may then have acommon switching line so that they are always activated simultaneously.However, they may alternatively have separate switching lines, so thatthe reset element and the discharge switching element can beindividually activated, for example for given modes of operation.Preferably, a plurality of sensor elements, for example all sensorelements of a row of the sensor matrix, have a common switching line forthe activation of the read-out switching elements. Such sensor elements,connected to a common switching line, can also have common switchinglines, or a common switching line for both elements, in order toactivate the reset elements or the discharge switching elements.

According to a particularly advantageous method of operating a sensoraccording to the invention the conversion element and the samplingcapacitor are reset to an initial state during a measuring and read-outcycle in each sensor element in a first phase. In a second phase avoltage difference which is representative of the conversion element inthe initial state is then adjusted across the sampling capacitor. Duringa third phase the voltage across the sampling capacitor is sustainedduring irradiation of the conversion element by means of a radiationsource whereas the voltage at the source output of the source followeris forced to change by the change of the signal at the conversionelement or of its capacitance. Evidently, in this context the termirradiation by means of the radiation source is to be understood to meannot only direct irradiation by the radiation source, but also indirectirradiation, for example after transmission through an object to beexamined. During a fourth phase the voltage difference across thesampling capacitor is adjusted to a value which is representative of theconversion element after the irradiation, the variation of the potentialat the side of the sampling capacitor which is connected to the read-outline then being measured as a measure of the radiation incident on theconversion element. Preferably, the variation of the charge at theread-out side of the sampling capacitor is then recorded in acharge-sensitive amplifier (CSA). This means that the amount of chargeflowing during the adjustment of the new voltage difference isintegrated.

As a result of this switching sequence a so-called “correlated doublesampling” (CDS) method is implemented in the relevant sensor element.This means that during the second phase a first sampling value isdetected for the conversion element in the stationary state whereasduring the fourth phase ultimately a value is measured across thesampling capacity which corresponds to the conversion element after theirradiation, only the difference between the initial state and theirradiated state being measured during the first sampling because of thebias in the second phase.

This switching process also offers the advantage that the resetoperation during the first phase lies outside the time interval in whichthe conversion element is irradiated and the signal is read out, so thatthe reset operation has no effect on the measuring result and hencecannot contribute to the noise.

According to a second version of the method of the invention, a darkcurrent is first detected on the conversion element during a firstsub-phase of the second phase, the voltage difference across thesampling capacitor being held during a given time interval withoutirradiation of the conversion element by the radiation source while atthe same time the voltage on the source output varies in conformity withthe dark current occurring across the conversion element. The darkcurrent can then be attributed essentially to leakage currents on theconversion element. This sub-phase is succeeded by a second sub-phaseduring which a voltage difference is adjusted across the samplingcapacitor, which voltage difference corresponds to a reference state ofthe conversion element after the detection of the dark current. Thesecond sub-phase is succeeded by a third sub-phase in which theconversion element is reset to its initial state, the voltage differenceacross the sampling capacitor then being maintained. The execution ofthe other phases is the same as in the previously described method.

The difference between this method and the previously mentioned mode ofoperation thus consists in that during the first sampling operation theinitial state, that is, the off-load voltage on the conversion element,is not taken as the reference value, but the reference state alreadycontains the integrated dark current. This means that the dark imagesare already buffered in the individual sensor elements and subtractedfrom the exposed images. Thus, the transfer and external storage of thedark images is dispensed with. Additionally, the usable dynamic range ofthe sensor is expanded, since the charges transferred from theindividual sensor elements no longer contain a dark current component.

The adjustment of the voltage difference across the sampling capacitorduring the second and the fourth phase is performed most easily byactivation of the read-out switching element, that is, via the read-outline. In order to sustain the voltage difference during the third phase,or during the dark current measurement, the read-out switching elementneed only be deactivated.

The resetting of the sampling capacitor in one embodiment of theinvention can be realized by activation of the discharge switchingelement connected parallel to the sampling capacitor, thus enablingaccelerated resetting.

The measuring and read-out cycles can be controlled in common for eachtime a plurality of sensor elements and via common switching lines. Thatis, after the irradiation the sensor elements in a sensor matrix aresuccessively read out in rows and are reset.

According to a third version of the method of the invention, there isprovided a method of operating a sensor having a plurality of sensorelements (10) arranged in rows and columns, each of which includes aradiation-sensitive conversion element (1) which generates an electricsignal in dependence on the incident radiation, a reset element (27)which resets the conversion element (1) to an initial state, and asource follower transistor (21) whose source is connected to an activeload (23) and to one side of a sampling capacitor (26) whose other sideis connected, via a read-out switching element (30), to a read-out line(8) for reading out the electric signal, the method comprising:

resetting the radiation-sensitive element and charging the samplingcapacitor of each pixel to a known voltage;

exposing the sensor to radiation, the radiation-sensitive conversionelement causing the voltage on the one side of the sampling capacitor tovary, wherein the read-out switching element is open during theexposure, providing an open circuit at the other side of the samplingcapacitor, thereby maintaining a constant charge on the samplingcapacitor; and

closing the read out switching elements and charging the samplingcapacitor for each pixel in a row to the voltage on the one side of thesampling capacitor, the amount of charge required being measured.

By separating the pixel resetting from the array readout phase, thisscheme provides sufficient time for the sampling capacitor to reach asteady state which eliminates the pixel offset error charge.Furthermore, the pixel readout time can be increased.

According to a fourth version of the method of the invention, there isprovided a method of operating a sensor having a plurality of sensorelements (10) arranged in rows and columns, each of which includes aradiation-sensitive conversion element (1) which generates an electricsignal in dependence on the incident radiation, a reset element (27)which resets the conversion element (1) to an initial state, and asource follower transistor (21) whose source is connected to an activeload (23) and to one side of a sampling capacitor (26) whose other sideis connected, via a read-out switching element (30), to a read-out line(8) for reading out the electric signal, the method comprising:

exposing the sensor to radiation with the read-out switching elementsclosed, the radiation-sensitive conversion element causing a change inthe voltage on the one side of the sampling capacitor and the read outline holding the other side of the sampling capacitor to a constantvoltage;

closing the reset elements to and opening the read out switchingelements, thereby holding the conversion element to a constant stateirrespective of the incident radiation;

closing the read out switching elements of each row in turn andmeasuring the charge stored on the sampling capacitors for each row inturn.

Since the reset switches remain on during the readout period, thephotodiode charge remains constant. Therefore, radiation incident on thedetector after the exposure time will not alter the signal being readoutthrough the sampling capacitors so that Frame Transfer operation ispossible.

An X-ray examination apparatus according to the invention includes anX-ray source for emitting an X-ray beam for irradiating an object so asto form an X-ray image, as well as a detector for forming an electricimage signal from said X-ray image, the X-ray detector being equippedwith a sensor according to the invention. Such an X-ray examinationapparatus has a particularly attractive signal-to-noise ratio and,therefore, is capable of operating with small doses, so that theradiation load for the object, notably a patient, can be kept small.

BRIEF DESCRIPTION OF THE DRAWINGS

Further details and advantages of the invention are disclosed in thedependent claims and the following description in which the embodimentsof the invention as shown in the Figures are described in detail. In theFigures:

FIG. 1 shows a circuit diagram of a sensor element of a sensor matrixaccording to the invention;

FIG. 2 shows the circuit diagram of FIG. 1 with the switching and supplylines leading tot he components in one embodiment;

FIG. 3 shows diagrammatically a switching sequence during a measuringand read-out cycle in conformity with a first mode of operation;

FIG. 4 shows diagrammatically a switching sequence during a measuringand read-out cycle in conformity with a second mode of operation;

FIG. 5 shows diagrammatically a switching sequence during a measuringand read-out cycle in conformity with a third mode of operation; and

FIG. 6 shows diagrammatically a switching sequence during a measuringand read-out cycle in conformity with a fourth mode of operation.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 1 shows a sensor element 10 in the form of a conventional FDXDmatrix cell 10 extended with the circuit according to the invention.Hundreds or thousands of such matrix cells 10 are arranged in rows andcolumns within a sensor.

Each matrix cell 10 includes first of all, as in customary FDXD matrixcells, a conversion element 1 with a storage capacitance 2 which may beintrinsically contained in the conversion element 1 or additionallybuilt in.

The conversion element 1 and the storage capacitance 2 are connected onone side to a counter electrode 9 which is common to all matrix cells10. Furthermore, each matrix cell 10 includes a read-out switchingtransistor 30 whose gate is connected to a switching line 7. Theswitching lines 7 are common to all matrix cells 10 of a matrix row. Theoutput of the switching transistors 30 is connected to a read-out line8, the matrix cells 10 of a column in the customary matrix detectorsbeing provided with a respective common read-out line 8.

The matrix cells 10 are row-wise activated for reading out, via theswitching lines 7, so that the individual matrix cells 10 of therelevant column are read out successively via each time the sameread-out line 8. At the end of the read-out line 8 there is provided acharge-sensitive amplifier (CSA) 11.

In the conventional FDXD matrix cells known thus far the conversionelement 1, or the storage capacitance 2, is connected directly to theinput of the switching transistor 30. This means that no amplificationtakes place within the individual matrix elements.

In the sensor according to the invention the side of the conversionelement 1, or the storage capacitance 2, which faces the counterelectrode 9 is connected first of all to the gate of a source followertransistor 21.

At the source output of the source follower transistor 21 there isprovided an additional transistor 23 which serves as an active load.Moreover, the output of the source follower transistor is connected to asampling capacitor 26, the other side of which is connected to the inputof the read-out switching transistor 30. The drain terminal 22 of thesource follower transistor 21 may be common to all matrix cells 10 andbe connected, for example, to the counter electrode 9. However, it mayalso be connected parallel to the switching line 7 so as to behorizontally common to all matrix cells 10 of a row.

Similarly, the gate terminal 24 and the source terminal 25 of the activeload 23 may be common to all matrix cells 10 of a sensor or common toone row only. It is in principle also possible to connect the gateterminal 24 directly to the drain of the active load 23.

The source follower transistor 21 as well as the active load 23 shouldpreferably operate in the saturation range of the relevant transistorcharacteristic, that is, the condition V_(DS)>V_(GS)−V_(t) must besatisfied, where V_(DS) is the drain source voltage, V_(GS) is the gatesource voltage and V_(t) is the actual threshold voltage of the relevanttransistor. The voltage transfer of the source follower transistor canthen be described by the equation V_(S)=V_(G)−V_(thr), where V_(thr) isthe effective threshold voltage which is dependent on the actualthreshold voltage V_(t) and the drain current I_(D). V_(S) is thevoltage present at the source and V_(G) is the voltage present at thegate of the source follower transistor 21.

A reset transistor 27 is connected to the output of the conversionelement 1, or the storage capacitance 2, which faces the counterelectrode 9; there is a reset transistor 27 which serves to bias theconversion element 1 and the parallel storage capacitance 2 to theoff-load voltage V_(GO). The source terminal 29 of the reset transistor27 can be constructed so as to be common to all matrix cells 10 of thesensor. It is also possible to form the output 29 for all matrix cells10 of a row, that is parallel to the switching line 7. The gate terminal28 of the reset transistor 27 is preferably constructed so as to becommon to all matrix cells 10 of a row.

Furthermore, the circuit also includes an optional dischargingtransistor 31 whose gate 32 is also connected, preferably via ahorizontal line, so as to be common all matrix cells 10 of a row.

FIG. 2 shows, by way of example, a circuit arrangement with fouradditional horizontal lines 3, 4, 5, 6, that is, lines which extendparallel to the switching line 7 and are common to all matrix cells 10of a matrix row. A switching line 3 is connected to the gate of theactive load 23. A further switching line 4 is connected to the gate 28of the reset transistor 27 and the gate 32 of the discharging transistor31. A third line 5 is connected to the source output 29 of the resettransistor 27 and a fourth line 6 is connected to the source output 25of the active load 23.

All components shown are integrated in the matrix cells 10 by way ofthin film technology. The transistors are made of amorphous silicon orpolycrystalline silicon.

Various preferred versions for operation of the proposed circuit will bedescribed in detail hereinafter. To this end, reference is made to therespective switching sequences which are diagrammatically shown in FIGS.3 to 6. The method shown in FIG. 4 utilizes double sampling while thatshown in FIG. 3 utilizes correlated double sampling (CDS) within therelevant matrix cell 10. FIG. 5 shows a different operating scheme whichensures the sampling capacitor is fully prepared before read out ofpixel data and FIG. 6 shows a method which enables Frame Transferdetector operation.

FIG. 3 shows the mode of operation called “switching noise suppression”.The top plot in FIG. 3 indicates the X-ray exposure time. The secondplot shows when the read out switching transistor 30 is on or off. Thethird plot shows when the reset transistor 27 and the dischargingtransistor 31 are on and off, and the bottom plot shows when theamplifier 11 is active. During a first phase I, that is, the resetphase, the reset transistor 27 is active in the matrix cells 10 of therelevant row. As a result, the conversion element 1 and the parallelstorage capacitance 2 of the magnitude C_(P) are biased to the off-loadvoltage. The voltage V_(G0) is then present at the gate of the sourcefollower transistor 21.

In as far as the circuit does not include the optional dischargingtransistor 31, the read-out switching transistor 30 remains activeduring the entire reset phase I (solid line). In the embodiment whichincludes the optional discharging transistor 31 as shown in FIG. 1 andFIG. 2, the discharging transistor 31 is active simultaneously with thereset transistor 27 in the reset phase I so as to achieve accelerateddischarging of the sampling capacitor 26 having the capacitance C_(S).The read-out switching transistor 30 is preferably deactivated duringthe reset phase I (dotted line).

After the end of the first phase I, the read-out switching transistor 30is active until an instant A in a second phase II. During this time thevoltage V_(G0)−V_(thr) is present at one side of the sampling capacitor26 whereas at the other side the input voltage V_(CSA) of the CSA 11arises via the lowpass filter formed by the sampling capacitor 26 andthe read-out switching transistor 30. The CSA 11 must be constructed insuch a manner that its input voltage is always maintained at theconstant value V_(CSA), irrespective of the fact whether the integratorin the CSA is active or not. Customary CSAs satisfy this requirement.Thus, the voltage difference V_(G0)−V_(thr)−V_(CSA) is maintained acrossthe sampling capacitor 26 as from the opening of the read-out switchingtransistor 30 at the instant A; this voltage difference isrepresentative of the reset conversion element 1. A “zero value” is thusquasi sampled.

The described reset operation in the first phase and the sampling of therelevant zero value in the second phase II are performed row-wise forall matrix cells 10 of the detector matrix. Subsequently, in theso-called X-ray window the entire detector matrix is exposed to X-raysduring the third phase III. The charge carrier pairs then generated inthe conversion element 1 of the relevant matrix cell 10 discharge thestorage capacitance 2 of the magnitude C_(P) by the signal charge Q_(P),with the result that the voltage at the gate of the source followertransistor 21 increases to V_(G1)=V_(G0)+(Q_(P)/C_(P)). The voltageV_(G1)−V_(thr) then arises at the output of the source followertransistor 21, without the voltage difference across the samplingcapacitor 26 being changed, because the read-out switching transistor 30and the discharging transistor 31 are inactive.

During the subsequent fourth phase IV of the row-wise red-out operation,first the integrator of the CSA 11 is activated and briefly thereafterthe read-out switching transistor 30 for each matrix row. Whereas theoutput of the source follower transistor 21 still carries the voltageV_(G1)−V_(thr), the other side of the sampling capacitor 26 carries theinput voltage of the CSA 11 again. At the instant B the integration inthe CSA 11 is stopped. The voltage difference across the samplingcapacitor 26 then amounts to V_(G1)−V_(thr)−V_(CSA). Comparison with thevoltage difference at the instant A reveals that the sampling capacitor26 has been subject to a change of charge amounting toQ_(S)=C_(S)*(V_(G1)−V_(G0))=C_(S)*Q_(P)/C_(P) during the integrationtime, that is, precisely only during this time. Therefore, exactly thischarge Q_(S) is measured as the result of the integration in the CSA 11.It is advantageous that the charge Q_(S) exceeds the change of the Q_(P)of the storage capacitance 2 by the charge amplification factorG_(Q)=C_(S)/C_(P). At the instant B an operation cycle of the matrixcell 10 is terminated and the described first phase I can commenceagain. This is shown in FIG. 3.

The described mode of operation is also compatible with a continuousX-ray exposure mode. For reasons of clarity, however, pulsed X-rayexposure was chosen. The leakage currents which practically always flowin the conversion elements 1 have also been ignored for the purpose ofsimplicity. As in conventional FDXDs, the leakage currents in this modeof operation are contained in the measured charge signal. Whenphotodiodes are used as the conversion element 1, it is to be noted thatthe capacitance C_(P) is not constant but dependent on the charge Q_(P),so that the transfer function contains a non-linear component.

The proposed solution has a particularly advantageous aspect which isformed by the stability of the transfer function of the circuit. Thisgain stability of the circuit is due to the fact that the sourcefollower transistor 21 has a stable voltage amplification amounting to 1which is converted into a charge amplification G_(Q)=C_(S)/C_(P) bymeans of the sampling capacitor 26. The offset stability is obtained bysubtraction of the relevant offset value from the overall valueconsisting of the signal and the offset value. As a result, all offseteffects which are slower in time than the image repetition time T_(F)are effectively eliminated. Due to the 1/F noise of the source followertransistors 21 and the active loads 23 used in the proposed circuit,additional noise may occur in this mode of operation. However, noisephenomena which are essentially slower than the image repetition rateT_(F) are again eliminated by the CDS method.

As regards the switching noise it is to be noted that the measuringresult is affected only by switching operations between and includingthe instants A and B. The reset operation for the conversion element 1and the sampling capacitor 26 does not lie within this time interval andhence does not contribute to the noise. The opening of the read-outswitching transistor 30 at the instant A makes a noise contributionwhich is a factor G_(Q) ^(1/2) larger than the switching noise of theknown FDXD. However, this is opposed by the signal amplification factorG_(Q), SO that overall the signal-to-noise ratio is improved by G_(Q)^(1/2). The switching noise upon deactivation of the integration in theCSA 11 at the instant B does not make an additional contribution,because it also occurs in conventional FDXD and in this case losessignificance in comparison with the signal because of the chargeamplification G_(Q).

Overall, this mode of operation leads to an enhanced signal-to-noiseratio when the charge amplification G_(Q) is sufficiently high; thisleads to a distinct enhancement of the image, notably in the case ofX-ray exposures with a low dose (for example, fluoroscopy).

The switching sequence for the second mode of operation, that is, theso-called dark current subtraction mode, is diagrammatically shown inFIG. 4. The top plot in FIG. 4 indicates the X-ray exposure time. Thesecond plot shows when the read out switching transistor 30 is on oroff. The third plot shows when the reset transistor 27 is on or off. Thefourth plot shows when the discharging transistor 31 is on and off, andthe bottom plot shows when the amplifier 11 is active. Like in the firstmode of operation, in the first phase I first the conversion element 1,or the capacitance 2, and the sampling capacitor 26 are biased to theoff-load voltage.

Subsequently, however, this value is not retained directly as the zerovalue on the sampling capacitor 26, but first a dark current is recordedon the conversion element 1 in a first sub-phase IIa. The dark currentof the relevant conversion element 1 is then integrated.

Subsequently, in a second sub-phase IIb a voltage difference is adjustedacross the sampling capacitor 26, that is, the dark image is sampled.This is realized by briefly activating of the read-out switchingtransistor 30.

After this first sampling in the sub-phase IIb, in the sub-phase IIc theconversion element 1 is reset, in this case neither the dischargingtransistor 31 nor the read-out switching transistor 30 is activated, sothat the voltage difference adjusted across the sampling capacitor 26during the first sampling operation is retained. When a dischargingtransistor 31 is used, of course, it must then be possible to switch thedischarging transistor 31 and the reset transistor 27 via separatelines.

After such second resetting of the conversion element 1, the X-rayexposure takes place in the X-ray window. Reading out in the phase IVtakes place like in the previously described “switching noisesuppression” mode of operation.

Overall, according to this method the measured value obtained at theinstant B after the integration in the CSA 11 is the chargeG_(S)=G_(Q)*(Q_(P)−Q_(D)) with the charge amplification factorG_(Q)=C_(S)/C_(P) as in the first mode of operation. The charge Q_(D)represents the dark current component integrated in the dark window.Therefore, the lengths of the dark window and the X-ray window, that is,the phases IIa and III, are preferably chosen to be the same, so that ameasured value which has been corrected in respect of the dark current.The quantities V_(thr) and V_(CSA) no longer occur in the measuredvalue, like in the previously described mode of operation, because theyare also eliminated by the subtraction.

Overall, the advantage of the second mode of operation resides in thefact that the dark images which are produced mainly by leakage currentsin the conversion elements 1 are subtracted already within theindividual matrix cells 10. Furthermore, this second mode of operationalso offers the advantages of a particularly advantageous stability ofthe transfer function of the circuit, that is, the gain stability andthe offset stability as already achieved for the first mode ofoperation.

Because resetting of the conversion element 1 and the storagecapacitance 2 takes place between the instants A and B within the thirdsub-phase IIc of the phase II, an additional noise component which isdue to the reset noise must be taken into account. Therefore, thesignal-to-noise ratio in the second mode of operation will be less thanthat in the first mode of operation.

The two modes described above each provide a reset operation of theconversion element 1 (phase I), followed by storage of a charge on thesampling capacitor 28 corresponding to the reset state of the conversionelement 1 (phase II). This is repeated for each row.

A potential problem with this approach is that the pixel must bereadout, reset and then the sampling capacitor 28 must be charged backto a steady state all within a short interval of about 20 μs. Assuming10 μs is required for readout, then this leaves 5 μs each for the othertwo tasks. Using typical device parameters for a-Si and poly-Si TFTsthere may be insufficient time for the sampling capacitor to be chargedback to a steady state within 5 μs. Ideally, a time period of 50-100 μsis appropriate.

If the sampling capacitor 28 is not recharged to a steady state, chargeoffsets from the pixel are caused.

FIG. 5 is used to illustrate an alternative timing scheme in which theresetting operation is not carried out during the read out of the array.Instead, all pixels in the array are reset (photodiodes reset andsampling capacitor charged to steady state) in parallel before the X-rayexposure. A 2 ms time interval is allocated for this so that thesampling capacitor can be easily charged to a steady state.

For illustration only, the timing sequence of FIG. 5 assumes 30 KHzoperation for a detector with a 1000×1000 array of pixels. This gives aframe time 50 of 33 ms. This may be divided in to an X-ray exposure timeof 13 ms and a line readout time of 20 μs.

The array readout is divided into three phases I, II and III, discussedbelow. The top plot in FIG. 5 shows the X-ray exposure time. The nextplot shows the read out pulse for the first row, and the next plot showsthe read out pulse for the last row. The fourth plot shows the state ofthe read out switching transistor 30 and the bottom plot shows the stateof the reset transistor 27.

Phase I

This is the 2 ms reset phase during which the sampling capacitor ischarged to the steady state value. At the start of the reset stage allof the reset transistors 27 are closed. The photodiode charge is resetand the gate of the source follower transistor 21 is fixed at the V_(GO)DC potential. The source of the source follower transistor 21 will reachthe steady state voltage V_(GO)−V_(thr). Next, the reset transistors 27open and the read out switching transistors 30 close. The voltage on oneplate (the top plate) of the sampling capacitor 26 is fixed at thesource voltage and other plate voltage is set to the column potential onthe read out line 8. Therefore, the charge on the sampling capacitor 26is constant at the end of the reset phase. Assuming that the columnpotential is OV, then the charge on capacitor 26 can be expressed as:

Q _(O) =C _(S) ×V _(SO)

where C_(S) is the capacitance of the sampling capacitor, and V_(SO) isthe initial (quiescent) value of the source voltage of the sourcefollower transistor 21 (equal to V_(GO)−V_(thr)).

Phase II

The signal exposure window follows the resetting stage. During thistime, photons incident onto the photodiode will generate a photocurrentto discharge the photodiode capacitance. This results in a linearincrease in the gate voltage. As described above, the active loadensures that the source voltage follows the gate voltage to maintain aconstant gate-source voltage. If the change in gate voltage during theexposure was ΔV_(pd), then the final value of source voltage V_(S1) willbe:

V _(S1) =V _(S0) +ΔV _(pd)

During the exposure time, the charge on capacitor 26 remains constantsince the read out switching transistor 30 is open

Phase III

The final stage in the readout sequence is the line by line readout.During readout, the read out switching transistors 30 of all pixels in arow are closed and the sampling capacitor 26 is charged to the new valueof the source voltage. The pulses may last 18 μs. The charge on thesampling capacitor at the end of the readout will be:

Q ₁ =C _(S) ×V _(S1) =C _(S)×(V _(S0) +ΔV _(pd))

During the readout period, the amplifier samples the change in chargeacross capacitor 26 as given below:

ΔQ _(C) =Q ₁ −Q ₀ =C _(S) ×ΔV _(pd)

This can be rewritten as:${\Delta \quad Q_{C}} = {\frac{C_{S}}{C_{P}} \times \Delta \quad Q_{pd}}$

where ΔQ_(pd) is the change in photodiode charge during an exposure, andthe term C_(S)/C_(P) is the pixel gain.

By separating the pixel resetting from the array readout phase, thisscheme provides sufficient time for the sampling capacitor to reach asteady state which eliminates the pixel offset error charge.Furthermore, the pixel readout time can almost be doubled from forexample 10 μs to 18 μs and this increases the time to read the pixelsignal.

FIG. 6 shows a further drive scheme that provides Frame TransferOperation. There is a desire to find a solution for operating X-raydetectors in bi-plane cardio applications. Such applications uses twodetectors orthogonally positioned and two X-ray sources operating atfast frame rates (60 Hz). The X-ray sources are pulsed sequentially sothat a first detector detects a dose from a first source and then thesecond detector detects a dose from the second source. However, some ofthe scattered dose from one source will be incident onto the detectorintended for the other source. Therefore when the first detector isbeing readout, the scattered X-rays from the second source will alterthe photodiode signal and hence image that is been read. Consequently itis necessary that during readout, the first detector is insensitive toX-rays from the second source and the second detector is insensitive toX-rays from the first source. This mode of detector operation is calledFrame Transfer. It literally means that during the exposure, the pixeldata is “stored/transferred” onto a storage device within the pixel.After the exposure, the storage device is read and any signal on thephotodiodes from scattered X-rays will not effect the signal beingreadout.

In the conventional X-ray detector, in which the pixels comprise a TFTand photodiode, the detector is exposed to an X-ray dose and then eachline is readout in turn. The readout process resets the pixels so thatat the end of the readout period, the array is reset. Therefore, afterthe exposure, the detector is still sensitive to X-rays until the entirearray has been read. For this reason, the standard detector does notprovide Frame Transfer operation.

A modified readout technique can provide the frame transfer operation,and the timing scheme is shown in FIG. 6. Again, the readout sequencecomprises an exposure (phase I) followed by a series of line readouts(phase II). The top plot in FIG. 6 shows the X-ray exposure time. Thesecond plot shows the state of all read out switching transistors 30 andthe next plot shows the state of all reset transistors 27. The next plotshows two read out pulses for the first row, and the last plot shows tworead out pulses for the last row.

In this timing arrangement, during the exposure (phase I), the change insource voltage of the source follower transistor 21 is transferredimmediately to the storage capacitor. For this purpose, all the read outswitching transistors are closed during exposure, so that the samplingcapacitors will be charged to the source voltage during exposure.

Assume that the gate voltage at the end of the exposure is:

V _(G1) =V _(G0) +ΔV _(pd)

and that the source voltage is:

V _(S1) =V _(bias) +ΔV _(pd)

where ΔV_(pd) is the change in source follower gate voltage during anexposure, V_(G0) is the DC voltage on the source follower gate at thestart of the exposure and V_(bias) is the quiescent voltage on thesource node. Therefore, the charge stored on the sampling capacitor is:

Q _(C) =C _(S) ×V _(S1) =C _(S)×(V _(bias) +ΔV _(pd))

After the exposure period, all of the read out switching transistors 30open and the reset transistors close. The source follower gate andsource nodes are reset to V_(G1) and V_(bias). Since the resettransistors 27 remain on during the readout period, the photodiodecharge remains constant. Therefore, X-rays incident on the detectorafter the exposure time will not alter the signal being readout throughthe sampling capacitors so that Frame Transfer operation is possible.

During the readout period, each read out switching transistor issequentially addressed. During this time, the sampling capacitor in anaddressed pixel will be charged to the quiescent source voltage(V_(bias)). Therefore, the change in charge during readout is:

ΔQ _(C) =C×ΔV _(pd)

=>ΔQ _(C) =C _(S) /C _(P) ×ΔQ _(pd)

and this charge is detected by the amplifier.

Thus, the pixel can be used to provide frame transfer mode of operationand maintain pixel gain with a gain of C_(S)/C_(P). Alternatively, thesampling capacitor could have the same value as C_(P) so there would beframe transfer without gain.

Finally, it is also to be noted again that all of said advantages areachieved by means of comparatively few additional components in theindividual sensor cells and without taking additional process stepsduring the production. Therefore, the manufacture of such sensorsaccording to the invention is hardly more expensive than the manufactureof sensors commercially available thus far.

What is claimed is:
 1. A sensor with a plurality of sensor elements (10), each of which includes a radiation-sensitive conversion element (1) which generates an electric signal in dependence on the incident radiation, and also with means (21 to 26) for amplifying the electric signal in each sensor element (10) and a read-out switching element (30) in each sensor element (10) which is connected to a read-out line (8) in order to read out the electric signal, characterized in that the means for amplifying include a respective source follower transistor (21) whose gate is connected to the conversion element (1), whose source is connected to an active load (23) and to one side of a sampling capacitor (26), the other side of the sampling capacitor (26) being connected to the read-out line (8) via the read-out switching element (30), and that a respective reset element (27) is connected to the conversion element (1) to reset the conversion element (1) to an initial state.
 2. A sensor as claimed in claim 1, characterized in that a discharge switching element is connected parallel to the sampling capacitor.
 3. A sensor as claimed in claim 1 or 2, characterized in that the active load and/or the read-out switching element and/or the reset element and/or the discharge switching element include transistors.
 4. A sensor as claimed in claim 2, characterized in that the reset element and the discharge switching element have a common switching line or separate switching lines for activating the relevant element.
 5. A sensor as claimed in claim 2, characterized in that a plurality of sensor elements have a common switching line for activating their read-out switching elements, and that these sensor elements also include common switching lines or a common switching line for activating their reset elements and/or their discharge switching elements.
 6. An X-ray examination apparatus, including an X-ray source for emitting an X-ray beam for irradiating an object so as to form an X-ray image, as well as a detector for generating an electric image signal from said X-ray image, characterized in that an X-ray detector includes a sensor as claimed in claim
 1. 7. A method of operating a sensor having a plurality of sensor elements, each of which includes a radiation-sensitive conversion element which generates an electric signal in dependence on the incident radiation, a reset element which resets the conversion element to an initial state, and a source follower transistor whose source is connected to an active load and to one side of a sampling capacitor whose other side is connected, via a read-out switching element, to a read-out line for reading out the electric signal, wherein: during a measuring and read-out cycle in each sensor element the conversion element and the sampling capacitor are reset to an initial state during a first phase, a voltage difference which is representative of the conversion element in the initial state is adjusted across the sampling capacitor during a second phase, the voltage across the sampling capacitor is sustained during a third phase while the conversion element is irradiated by means of a radiation source, and during a fourth phase the voltage difference across the sampling capacitor is adjusted to a value which is representative of the conversion element after the irradiation, the variation of the potential at the side of the sampling capacitor which is connected to the read-out line being measured as a measure of the radiation incident on the conversion element.
 8. A method as claimed in claim 7, wherein within the second phase first a dark current is recorded on the conversion element in a first sub-phase, and that subsequently there is a second sub-phase in which a voltage difference is adjusted across the sampling capacitor, which voltage difference corresponds to a reference state of the conversion element after the recording of the dark current, and that subsequently in a third sub-phase the conversion element is reset to its initial state while the voltage difference across the sampling capacitor is maintained.
 9. A method as claimed in claim 7 or 8, wherein the adjustment of the voltage difference across the sampling capacitor in the second and the fourth phase takes place by activation of the read-out switching element, and that the read-out switching element is deactivated in order to sustain the voltage difference.
 10. A method as claimed in claim 7, wherein the sampling capacitor is reset by activation of a discharge switching element connected parallel to the sampling capacitor.
 11. A method as claimed in claim 7, wherein a measuring and read-out cycle is controlled for a plurality of sensor elements in common via common switching lines.
 12. A method of operating a sensor having a plurality of sensor elements arranged in rows and columns, each of which includes a radiation-sensitive conversion element which generates an electric signal in dependence on the incident radiation, a reset element which resets the conversion element to an initial state, and a source follower transistor whose source is connected to an active load and to one side of a sampling capacitor whose other side is connected, via a read-out switching element, to a read-out line for reading out the electric signal, the method comprising: resetting the radiation-sensitive element and charging the sampling capacitor of each pixel to a known voltage; exposing the sensor to radiation, the radiation-sensitive conversion element causing the voltage on the one side of the sampling capacitor to vary, wherein the read-out switching element is open during the exposure, providing an open circuit at the other side of the sampling capacitor, thereby maintaining a constant charge on the sampling capacitor; and closing the read out switching elements and charging the sampling capacitor for each pixel in a row to the voltage on the one side of the sampling capacitor, the amount of charge required being measured.
 13. A method of operating a sensor having a plurality of sensor elements arranged in rows and columns, each of which includes a radiation-sensitive conversion element which generates an electric signal in dependence on the incident radiation, a reset element which resets the conversion element to an initial state, and a source follower transistor whose source is connected to an active load and to one side of a sampling capacitor whose other side is connected, via a read-out switching element, to a read-out line for reading out the electric signal, the method comprising: exposing the sensor to radiation with the read-out switching elements closed, the radiation-sensitive conversion element causing a change in the voltage on the one side of the sampling capacitor and the read out line holding the other side of the sampling capacitor to a constant voltage; closing the reset elements to and opening the read out switching elements, thereby holding the conversion element to a constant state irrespective of the incident radiation; closing the read out switching elements of each row in turn and measuring the charge stored on the sampling capacitors for each row in turn. 